AN1137 using the MCP1631 family to develop low cost battery chargers

AN1137 Using the MCP1631 Family to Develop Low-Cost Battery Chargers Author: Terry Cleveland Microchip Technology Inc INTRODUCTION As portable rechargable applications continue to grow, there is an increase in demand for unique or custom battery charger designs In addition to the increase in portable rechargable applications, battery chemistry continues to improve and with that new charge methods and profiles are emerging This all leads to the increase in demand for new or custom charge profile designs In this application note, a mixed signal multichemistry battery charger design technique will be discussed that can accommodate the changing portable power management world The reliability and safety concerns with charging batteries can also benefit from programmable mixed signal designs Charge rates and constant voltage levels can be updated in the field with a change in firmware This allows the user to adapt to new smart battery packs and select desired runtime versus cycle life By charging the battery to a lower constant voltage, the run time is shortened but the number of charge cycles will increase Another programmable battery charger feature is its ability to charge multi-chemistry battery packs By detecting the number of cells and cell chemistry, a programmable charger can adapt to a new battery pack This enables customers to choose between portability, runtime and cost when purchasing a portable system © 2007 Microchip Technology Inc COMMON CHARGE PROFILES NiMH Charge Profile Figure shows a typical charge profile for NiMH batteries The charge cycle begins once a battery is detected by regulating a small current or conditioning current into the battery pack If the cell voltage is above 0.9V per cell, it is safe to charge the pack with a fast charge or high current (for NiMH or NiCd, this current can range from 50% to over 100% of the batteries capacity) When the battery reaches capacity, cell manufactures recommend a top-off charge to complete the charge cycle It is typically not recommended to trickle charge NiMH batteries, this can lead to overheating and reduced battery life Fast charge termination for NiMH batteries can be tricky As the battery reaches capacity, it no longer can accept a charge The energy from the charger that was stored in the battery, now turns into heat causing the battery temperature to rise There are two primary methods to determine when the battery has reached full charge, one is a sudden increase in temperature, the other being a subtle drop in battery voltage or -dV/dt With NiMH batteries, the -dV/dt can be difficult to detect, since the change can be very small, especially with lower charge rate designs The +dT/dt or temperature rise is typically easier to detect For a robust design, both methods should be used so either can terminate the fast charge portion of the charge cycle Once the fast charge is terminated, a timed top off charge is recommended, a continuous constant charge is not recommended for NiMH batteries DS01137A-page AN1137 NiMH and NiCd Charge Profile Stage Pre-Charge Stage Fast Charge Stage End Fast Charge Stage Top Off Charge -dV/dt 0.8V VCELL ICH = 1.0C -dT/dt VCELL (V) ICH (A) T(°C) FIGURE 1: ICH = 0.2C Pack T (°C) hour ICH = 0.05C NiMH / NiCd Charge Profile Li-Ion Charge Profile The charge profile for Li-Ion batteries starts with cell qualification The cell voltage should be greater than 3.0V per cell before initiating a fast or high current charge If the cell voltage is less than 3.0V per cell, a low value conditioning current is used to start the charge cycle Once the cell voltage is above the 3.0V threshold, a fast charge or high current charge is initiated (0.5C to 1.0C) As the battery cell voltage rises, it reaches the maximum voltage value before it reaches full capacity As an example, most Li-Ion batteries constant voltage level is 4.2V, where the battery charger now transitions into a constant voltage source (regulating voltage instead of current) The charge cycle continues as the charge current decreases while in the constant voltage mode Once the charge current decreases to about 7% of the fast charge value, charge is terminated Continuing the charge cycle past this point can damage the battery so the charge must be terminated Once terminated a new charge cycle can be initiated when the battery voltage decreases to approximately 4.0V DS01137A-page © 2007 Microchip Technology Inc AN1137 Li-Ion Charge Profile Stage Pre-Charge Stage Constant Current VCELL FIGURE 2: ICH = 0.2C Stage Termination 4.2V ICH = 1.0C 2.8V VCELL (V) ICH (A) T(°C) Stage Constant Voltage ICH = 0.07C Pack T (°C) ICH = 0C Li-Ion Charge Profile Multi-Chemistry Charger There are significant differences in the charge profile between Ni batteries versus Li-Ion batteries A multichemistry charger must be able to implement the proper profile and proper termination methods This application note will demonstrate a charger that has the capability to charge single or multiple cells in series THE POWER BEHIND CHARGING BATTERIES · – Eff P DISS = P OUT × ⎛ ⎞ ⎝ Eff ⎠ A switching charger solution operating at similar conditions at 85% efficiency would dissipate approximately 1.05 Watts, making it much easier to cool For high input voltage applications, switching battery chargers are smaller and more cost effective A battery charger and power supply have a lot in common, delivering a regulated output from a varying input Two solutions are prevalent, linear and switch mode solutions The linear solution is commonly used for low input voltage or low power applications Its main drawback is internal power dissipation, calculated by the following formula: P DISS = ( V IN – V BATT ) × I CHARGE For example, a +12V input linear charger would dissipate 18 watts when charging a +3.0V Li-Ion battery at 2A Any power dissipation over a few watts is a challenge to cool Cooling 18 watts of power dissipation is no easy task, airflow and large heatsinks are required making a linear solution impractical © 2007 Microchip Technology Inc DS01137A-page AN1137 CHARGER POWER TOPOLOGY • Primary Inductive Converter: - The SEPIC converter topology has an inductor at the input, smoothing input current reducing necessary filtering and generated source noise • Single Low Side Switch: - A single low side switch reduces MOSFET drive and current limit protection complexity • Buck-Boost Capability: - For applications where the input voltage can be above or below the battery voltage a SEPIC can buck or boost the input voltage Many switching regulator power topologies exist, buck, boost, SEPIC and flyback are all used to develop switching battery chargers (including others for very high power applications) A SEPIC converter is commonly used, it has advantages over buck and boost converters when used in battery charger applications • Capacitive Isolation: - There is no direct dc path from input to output providing isolation, this results in less power components and a safer battery charger SEPIC Converter Capacitive Isolation Coupled Inductor Blocking Diode CC +12V Input +Vbatt CIN COUT ISENSE Switch Batteries VEXT CS Input Current FIGURE 3: SEPIC Topology MULTI-CHEMISTRY BATTERY CHARGER DESIGN The development of an intelligent multi-chemistry battery charger starts with the microcontroller By implementing the charge algorithm in code, the charger can be adapted for multi-chemistry, custom charge profile and unique applications For dc-dc converters, switching at high frequency with high performance gate drive capability, PWM control and high-speed protection, specialized analog circuitry is required A new high-speed analog PWM, the MCP1631HV was developed for constant current SEPIC applications (battery chargers and LED drivers) By implementing the pulse width modulation, PWM, control using the DS01137A-page RLIMIT MCP1631, the battery charger has the benefits of analog speed and resolution By controlling the charge algorithm using the microcontroller, the battery charger has the intelligence and flexibility to generate a profile for all battery types using digital timers and programmed algorithms As complex as this project sounds, it is really quite simple if the SEPIC converter is thought of as a microcontroller controlled current source To increase current, the microcontroller simply increases the VREF input to the MCP1631HV and to decrease current, the microcontroller decreases the VREF input to the MCP1631HV To generate a charge algorithm, the microcontroller measures the battery voltage using an © 2007 Microchip Technology Inc AN1137 analog to digital converter(A/D), computes the desired charge current and adjusts the SEPIC controlled current source up or down To develop the charge algorithm for the NiMH battery, the microcontroller A/D converter is used to measure the battery pack voltage, when the pack voltage is within the desired range, the microcontroller sets the proper current level To terminate the charge, two A/D inputs are used, one to sense the decreasing battery voltage and one to sense the increasing battery pack temperature Charge termination will occur, if either one or both are detected To develop the algorithm for charging Li-Ion batteries, the A/D converter is used to measure pack voltage Depending on pack voltage, the microcontroller will set the appropriate charge current Once the pack voltage reaches the constant voltage phase, the A/D converter senses and regulates the pack voltage by adjusting the amount of current into the battery The current continues to decrease until is reaches about 7% of the fast charge value At this point, the microcontroller terminates the charge VIN = +5.3V to +16.0V AVDD_OUT = +5.0V C C 250 mA Available MCP1631HV µController VDD Input VIN = +3.8V to +16.0V AVDD_OUT = +3.3V C C 250 mA Available MCP1631HV µController VDD Input The MCP1631HV Implementation The MCP1631HV integrates the necessary blocks to develop an intelligent, programmable battery charger or constant current source used for driving high power LED’s INPUT VOLTAGE AND BIAS GENERATION The MCP1631HV provides a regulated bias voltage for internal circuitry that is available for biasing the microcontroller and other components It is available in two regulated voltage options, +5.0V and +3.3V and can handle a maximum output current of 250 mA The maximum input voltage range for the regulator is +16.0V and can withstand transients to +18.0V For regulated input voltages or higher input voltage applications, the MCP1631 device option without internal regulator can be used By using a high voltage regulator to bias the MCP1631 and microcontroller, the range of input voltage for the design is only limited by the regulator maximum input and power train design AVDD_IN = USB +5.0V C MCP1631 µController VDD Input High Voltage Input High Voltage Linear C Regulator Regulated +3.3V or +5.0V MCP1631 C µController FIGURE 4: MCP1631HV and MCP1631 Bias Voltage Options © 2007 Microchip Technology Inc DS01137A-page AN1137 HIGH SPEED ANALOG PWM OPERATION ramping current is used for peak current mode control CS signal A filter is used on the CS input to remove the leading edge turn on spike associated with the turn on of the external power MOSFET The driver P-Channel MOSFET is powered using a separate PVDD pin helping to keep switching noise off of the AVDD pin and sensitive CS circuitry The high-speed analog PWM is used to control the power train switch ON and OFF times to regulate the output of the converter Voltage or current can be regulated depending on what is being sensed For the SEPIC Battery Charger application, the MCP1631HV is always regulating current, the microcontroller is programming this current The error amplifier is configured as an integrator, so any difference between its inputs, VREF and VFB are quickly removed If the VFB input is high, the inverting error amplifiers output, (COMP), will be pulled down, lowering the peak current into the switch and lowering duty cycle bringing the output back into regulation The external R and C used for compensation is used to control the speed of the error amplifiers output response If not compensated properly, the error amplifier output will move to fast (unstable system with under damped oscillations) or slow (over damped system with no performance or response to changes) The VREF input is set by the microcontroller to program the proper charge current The analog PWM starts with the oscillator input, typically a microcontroller PWM output or simple clock output (50% duty cycle) When the oscillator input is high, the VEXT output is pulled low, (N-Channel MOSFET Driver is ON) A new cycle is started when the OSC_IN input transitions from a high to a low, the internal N-channel MOSFET driver turns off and the PChannel MOSFET turns on driving the VEXT pin high turning on the external N-Channel MOSFET Current begins to ramp up in the external CS sense resistor until it reaches 1/3 of the level of the error amplifier output voltage (limited to 0.9V by error amplifier clamp) The 0.9V limit is used as an overcurrent limit, the OSC_IN Low = Active Duty Cycle High = VEXT OFF V = COMP/3 PVDD Latch S MOSFET P CS INPUT VEXT Q VREF VFB + A1 + C1 - COMP 2R - R PGND R Error Amplifier and Compensation MOSFET N High Speed Comparator Note 1: A1 output or COMP is clamped to 2.7V maximum to set current limit FIGURE 5: DS01137A-page Analog PWM Operation © 2007 Microchip Technology Inc AN1137 CURRENT REGULATION MCP1631HV integrates an inverting 10V/V gain amplifier to increase the battery current sense signal The microcontroller sets the VREF input to the desired current level, the MCP1631HV uses the VREF input as a reference for regulation To sense battery current for regulation in a SEPIC converter, the secondary winding of the coupled inductor can be used The average current flowing through the secondary winding is equal to the current flowing into the battery As shown, this topology does not require the sense resistor in series with the battery, removing any power lost in series with the battery while running the system When sensing battery current, a low value sense resistor is desired to minimize power loss, the The resistor in series with the external SEPIC switch provides a high speed current limit protecting the switch and other power train components from a short circuit or over current condition VIN CIN CS INPUT VREF VFB + COUT + C1 - 2R A1 - IBATT R BATT IINPUT 10R 10X IBATT FIGURE 6: R A2 + R Current Regulation Diagram SENSING BATTERY VOLTAGE Using the internal microcontroller A/D converter to sense battery voltage is a popular approach An issue with this technique is the A/D converter requires a low source impedance to perform accurate readings Low source impedance requires low resistance values that draw excessive quiescent current from the battery The MCP1631HV integrates a low current amplifier (A3), configured as a unity gain buffer The buffer output impedance is low, driving the SAR A/D converter, while consuming very little quiescent current A high value resistor divider is used to drop the battery voltage to an acceptable range R1, R2 and R3 values are selected to minimize the drain on the batteries, typically drawing on the order of µA The microcontroller reads the A/D converter, calculates the current setting and adjusts the VREF input to regulate current removed or opens OV protection is typically required for any current source application (battery chargers, LED drivers) The MCP1631HV integrates an internal high speed OV comparator that has a 1.2V reference connected to its inverting input If the voltage on the OV_IN pin exceeds the 1.2V threshold, the VEXT output is asychronously terminated Switching will resume after the voltage has dropped more than the built in 50 mV of hysteresis If a battery is removed during the charge cycle, the charger output voltage will be limited to a safe value Overvoltage (OV) protection is a common battery charger protection feature The OV protection is not there to protect the battery, it is used to protect the power train from excessive voltage if the battery is © 2007 Microchip Technology Inc DS01137A-page AN1137 CS INPUT COUT VREF VFB + + C1 - 2R A1 - To PWM Latch H = PWM OFF BATT R R1 VS_OUT to microcontroller A/D Converter + A3 - R2 OV COMP + C2 - R3 +1.2V Comp FIGURE 7: DS01137A-page MCP1631HV Voltage Buffer and Overvoltage Comparator Setup © 2007 Microchip Technology Inc AN1137 System Level Block Diagram The system level block diagram shown in Figure represents all of the MCP1631HV internal blocks The SHDN input is used to turn off the charger and lower the quiescent current draw to a 4.4 µA typical, the +5V generated bias is available and A3 remain powered for battery monitoring and microcontroller power MCP1631HV/VHV High-Speed Analog PWM +3.3V or +5.0V LDO 250 mA VIN VDD Internal 1.2V VREF VDD AVDD_OUT / AVDD_IN Shutdown Control A3 Remains On SHDN Overvoltage Comp w/ Hysteresis C2 + OVIN PVDD OSCDIS VDD 100 kΩ 0.1 µA OT OSCIN VEXT UVLO S VDD PGND Q VDD + C1 - COMP R A2 + A1 + 2R ISIN R VDD 2.7V Clamp Note 1: For Shutdown control, amplifier A3 remains functional so battery voltage can be sensed during discharge phase A3 VSIN - R AGND R ISOUT + VREF 100 kΩ 10R VDD VDD FB Q - CS/VRAMP VSOUT FIGURE 8: MCP1631HV Block Diagram © 2007 Microchip Technology Inc DS01137A-page AN1137 Charger Reference Board Design charger application A battery charger reference design was developed for the MCP1631HV to evaluate the device in a battery Multi-cell, Multi-Chemistry Charger VIN Range +5.5V to +16V L1A CC SCHOTTKY DIODE COUT CIN L1B MCP1631HV +VDD_OUT VEXT VIN ILIMIT BATTERY ISENSE CS AVDD_OUT FSW SET RTHERM PGND PVDD OSCIN ISIN ISOUT OVIN NC VSIN FB VREF NC SHDN COMP OSCDIS AGND VSOUT 0V PROTECTION LOW IQ SHUTDOWN PROGRAMMAGLE CURRENT SOURCE REFERENCE R C PIC® Microcontroller VDD GP1/C CCP1 GP3 GP4 GP5 GND GP0/C AVDD_OUT LED STATUS INDICATOR FIGURE 9: DS01137A-page 10 Charger Diagram © 2007 Microchip Technology Inc AN1137 A K A K A K A K A K A K K A 28 27 26 25 24 23 22 21 20 19 18 17 16 15 10 11 12 13 14 K A FIGURE 10: Detailed Schematic © 2007 Microchip Technology Inc DS01137A-page 11 AN1137 FIGURE 11: DS01137A-page 12 Board Layout © 2007 Microchip Technology Inc AN1137 THE DESIGN DETAILS OF CHARGING BATTERIES USING THE PIC® MICROCONTROLLER AND MCP1631HV WITH A SEPIC TOPOLOGY Design Example: • • • • • • VIN = 12V VBATT = 0V to 4.2V IBATT = 200 mA Pre-Charge Current IBATT = 2A Fast Charge Current IBATT = 140 mA termination or “tail” current Overvoltage Protection SEPIC Power Train Design • Calculate Maximum Output Power P OUT = V BATT × I BATT POUT = 4.2V X 2.0A or 8.4 Watts P OUT P IN = -Efficiency • By making an efficiency estimate, the converter input power can be estimated The typical efficiency of a SEPIC converter in this power range using a schottky diode for the output rectifier is around 85% • PIN = 8.4 Watts / 0.85 or 9.88 Watts • IIN = PIN / IIN - IIN = +12V / 0.88 Watts - IIN = 1.21 A With IIN and IBATT known, the average inductor current for each winding is known Inductor Ripple Current For the coupled inductor, the effective inductance is twice the value of the inductor, this is a result of 2x the voltage across 2x the number of turns Since the value of L is proportional to n2, the effective inductance is twice the actual value of the inductor 2×V -2 n © 2007 Microchip Technology Inc A 10 µH inductor looks like a 20 µH inductor (for coupled inductors only) Larger inductance reduces ripple current and operates in the continuous mode at lighter loads, an advantage over non-coupled inductor solutions The input and output inductor ripple current is equal to: VL ΔI L = × t ON L Where TON is the amount of time the SEPIC switch is turned on: t ON = DutyCycle × -F SW Where Duty Cycle for a SEPIC converter operating in continuous conduction mode is equal to: V OUT DutyCycle = V OUT + V IN To derive the transfer function of the SEPIC converter, start by balancing the inductor volt-time product in the boost stage (W1) Q1 Turned on (+ Slope): ΔI W1 ⁄ t ON = V IN ⁄ L W1 Q1 Turned off (- Slope): V C1 + V OUT – V IN ΔI W1 ⁄ t OFF = -L W1 Inductor slope’s must be equal for volt-time balance: V IN V C1 + V OUT – V IN t ON × = t OFF × -L W1 L W1 Multiply both sides by 1/(tON + tOFF): V IN × D = ( V C1 + V OUT – V IN ) × ( – D ) Solve for VC1: V C1 = V IN × ⎛ -⎞ – V OUT ⎝ – D⎠ For the second stage, the inductor slopes must also be equal Q1 Turned on (+ Slope): ΔI W2 V C1 = t ON L W2 DS01137A-page 13 AN1137 Q1 Turned off (- Slope): V OUT ΔI W2 = t OFF L W2 Inductor slope’s must me equal for volt-time balance: V C1 V OUT t ON × = t OFF × L W2 L W2 Multiply both sides by 1/(tON + tOFF): V C1 × D = V OUT × ( – D ) Solving for VC1 1–D V C1 = V OUT × ⎛⎝ -⎞⎠ D Set VC1 = VC1 for both the Boost stage and the BuckBoost stage: 1–D V C1 = V IN × ⎛⎝ -⎞⎠ – V OUT = V OUT × ⎛⎝ -⎞⎠ 1–D D Solving for VOUT/VIN: V OUT D - = ⎛ -⎞ ⎝ – D⎠ V IN Looking back, if D/(1-D) x VIN is substituted for VOUT, it is shown that VC1 = VIN This is true, if C1 is large enough that the ripple voltage on C1 is low Now that the duty cycle is known as a VOUT/VIN relationship, the duty cycle can be calculated for any input output condition Remember, this transfer function is dependent upon the fact that inductor current is continuous or never reached zero If it does reach zero, this transfer function is no longer true and there is another state added to the operation DS01137A-page 14 © 2007 Microchip Technology Inc AN1137 Power Train Design W1 IW1 + VIN C1 + D Q1 IW2 W2 COUT + ICHARGE A W1 + IW1 + VIN C1 IW2 W2 COUT + ICHARGE B W1 IW1 + VIN C + IW2 W2 COUT + ICHARGE C Sum of Winding Currents Diode Current IW2 Switch Current IW1 + IW2 tON tOFF IOUT (Average) IOUT x (VOUT/VIN) IIN (AVERAGE) IW1 tON FIGURE 12: tOFF tON tOFF SEPIC Converter Inductor, Switch and Diode Currents © 2007 Microchip Technology Inc DS01137A-page 15 AN1137 Inductor Winding Current Calculation The first step to calculating the inductor winding current is to know the maximum output power For this constant current battery charger application, the output power is simply the maximum output voltage times the charge current P OUT = V OUT × I CHARGE Maximum output voltage is equal to 4.2V (1 battery @ 4.2V) POUT =4.2V x A or 8.4 Watts Since energy is conserved, the input power is equal to the output power (assuming 100% efficiency) An efficiency estimate can be used to closer approximate the input current P IN = P OUT ⁄ ( Efficiency ) Where: PIN PIN = 8.4 Watts / 85%; 85% used as a typical efficiency estimate = 9.88 Watts The average input current is equal to the input power divided by the input voltage: I IN ( AVG ) ) = P IN ⁄ V IN Where: IINAVG = 9.88 Watts/12V (Nominal) IINAVG = 824 mA (Typical average input current The peak-to-peak W1 inductor current ripple calculation was shown earlier Given the derived transfer function and the maximum voltage on the output of the converter to be 4.2V, the switch on time is estimated Switch On Time: t ON V OUT ⁄ ( V OUT + V IN ) = F SW For the 12V input and 4.2V output case, the switch ON time is estimated to be approximately 519 ns (500 kHz switching frequency) The input peak-to-peak ripple calculated: GIVEN: current can be LW1 = LW2 = 20 µH (10 µH Coupled) Input Peak-to-Peak Ripple Current (W1) ΔIL(W1) = (12V / 20 µH) x tON = 311 mA IL(W1)PK = IINAVG +1/2 x ΔIL(W1) IL(W1)PK = 980 mA for winding (W1) IL(W1)MIN = IINAVG -1/2 x ΔIL(W1) IL(W1)MIN = 669 mA for winding (W1) The ripple current in winding (W2) is calculated in a similar fashion The main difference is that the average current in W2 is equal to IOUT or 2A in this application W2 Peak-to-Peak Ripple Current ΔIL(W2) = (12V / 20 µH) x tON = 311 mA IL(W2)PK = IOUTAVG +1/2 x ΔIL(W2) IL(W2)PK = 2.16 A for winding (W2) IL(W2)MIN = IOUTAVG -1/2 x ΔIL(W2) IL(W2)MIN = 1.85 A for winding (W2) Note: In the case of VIN = VOUT, the current in W1 = W2 (ripple and average) The coupled inductor winding currents calculated above are used to determine the size of the inductor necessary High switching frequency has several advantages, smaller ripple current, lower peak and RMS current and lower volt-time product on the inductor core This leads to a small, low-cost solution SEPIC Switch Current and Voltage Calculations The switch current (IQ1) is equal to the combination of the winding currents during the switch on time When the switch is turned on, it conducts the current in W1 and W2 ISW = IW1 + IW2 = 2.82A (Average) ISWPK = 2.82A + 311 mA = 3.14A The minimum switch current is equal to: ISWMIN = 2.82A - 311 mA = 2.51A RMS of a Trapezoidal waveform I SWRMS = ⎛ I A + I A × I B + I B⎞ D × ⎜ -⎟ ⎝ ⎠ Where: IA = 2.51A = Minimum, IB = 3.14A = Maximum The RMS value of the switch current is approximately 1.44 mA DS01137A-page 16 © 2007 Microchip Technology Inc AN1137 The peak switch voltage is equal to VIN + VOUT for the SEPIC converter Any leakage inductance voltage spike is clamped through the output diode by the output capacitor A switch voltage rating for this application should be a minimum of VIN(MAX) + VOUT(MAX) VSW = 12V +4.2V VSW = 16.2V A 30V, 30 milli-ohm, logic-level switch is selected MOSFET switching losses should also be considered when selecting the MOSFET switch Low on resistance switches tend to have high capacitance and will switch slower, increasing switching losses The lowest RDSON MOSFET is not necessarily the best choice When using the SOIC-8 package for a 30V MOSFET, there are many choices available SEPIC Diode Voltage and Current Calculations A schottky diode is recommended for low-voltage applications For battery charger applications, the SEPIC diode will block current flow from the battery back to the input The reverse leakage current of the selected schottky diode can be a critical parameter, if low battery drain is desired Low schottky diode forward drop is also a key parameter; the low drop improves converter efficiency The maximum reverse voltage across the SEPIC diode occurs during the switch on time The cathode of the schottky diode is connected to VOUT, the anode of the schottky diode is connected to the SEPIC coupling capacitor The voltage across the coupling capacitor voltage is equal to VIN; the voltage across the diode is equal to VOUT - (-VIN) or VOUT + VIN The peak SEPIC diode current occurs when the switch is turned off The peak diode current is equal to the peak current in W2, plus the peak current in W1 or 3.14A The average diode current is equal to the output current (IOUT), typical of all topologies with a series diode in the path of the output SEPIC Coupling Capacitor (C1) RMS Current Calculations and Voltage Rating The RMS current in the SEPIC coupling capacitor is mainly dependant upon output power with some influence by inductor ripple current As output power increases, the capacitor ripple current will increase as well As shown in Figure 12 (during the switch on time), the current in winding (output current) is flowing through the coupling capacitor C1 During the switch off time, the C1 current is equal to the current in winding number (W1) As previously discussed, the W1 current is equal to the average input current Therefore, the worst case or maximum RMS current in the coupling capacitor will occur at maximum output power © 2007 Microchip Technology Inc and minimum input voltage To estimate size for the coupling capacitor, the capacitor derivative equation can be used dV I C = C × dt The rate of change of voltage across the capacitor is related to the amount of current through the capacitor and the size or energy storage capability of the capacitor For the SEPIC converter coupling capacitor, the voltage is approximated to be a DC value when deriving the duty cycle The ripple voltage should be no more than 5% of the voltage across the capacitor or the input voltage In this example, the input voltage and C1 DC voltage is 12V, so there should be no more than 5% or 600 mV of ripple on the coupling capacitor In this example there is an average of 2A flowing through the coupling capacitor during the switch on time The on time is approximately 26% or 520 ns To keep the capacitor voltage ripple less than 5% of VIN, or 600 mV, the amount of capacitance is equal to (2A) / (600mV/520 ns) or 1.73 µF For this application a standard value 2.2 µF X7R 25V rated ceramic capacitor should be used +IW2 IW2 IW1 IC1 - IW1 Note: Area above equals the area below zero FIGURE 13: C1 Ripple Current As shown in Figure 13, the coupling capacitor ripple current is largely dependent upon output power and input voltage As the input voltage decreases, the current in W1 increases During the switch on time, the current flowing in W2 is equal to the current flowing in C1 When the switch turns off, the current quickly changes magnitude and direction so that the current flowing in C1 is equal to the current in W1, magnitude and direction DS01137A-page 17 AN1137 As an approximation, the RMS current in C1: I C1 ( RMS ) = I OUT × V OUT ⁄ V IN For worst-case situations, the RMS current in the C1 coupling capacitor is equal to 2A x (4.2V / 12V)1/2 or 1.18A The current rating for small multi-layer ceramic capacitors is typically much higher than 1.18A For higher power applications, it may be necessary to use multiple capacitors in parallel to keep the RMS current within ratings DS01137A-page 18 CONCLUSION For applications that require intelligent power management solutions like battery chargers, the combination of a microcontroller and MCP1631 high-speed PWM is very powerful It brings the programmability benefits of the microcontroller and adds the performance of a high-speed analog PWM The analog PWM will respond to changes in input voltage and output current very quickly No code or execution time is necessary to regulate or protect the circuit The microcontroller is used for programmability, establishing charge profile conditions and monitoring the circuit for fault conditions and taking the appropriate action, in the event of a specific fault © 2007 Microchip Technology Inc Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions • There are dishonest and possibly illegal methods used to breach the code protection feature All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets Most likely, the person doing so is engaged in theft of intellectual property • Microchip is willing to work with the customer who is concerned about the integrity of their code • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving We at Microchip are committed to continuously improving the code protection features of our products Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates It is your responsibility to ensure that your application meets with your specifications MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE Microchip disclaims all liability arising from this information and its use Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, KEELOQ logo, microID, MPLAB, PIC, PICmicro, PICSTART, PRO MATE, rfPIC and SmartShunt are registered trademarks of Microchip Technology Incorporated in the U.S.A and other countries AmpLab, FilterLab, Linear Active Thermistor, Migratable Memory, MXDEV, MXLAB, SEEVAL, SmartSensor and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP, ICEPIC, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, PICkit, PICDEM, PICDEM.net, PICLAB, PICtail, PowerCal, PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, Select Mode, Smart Serial, SmartTel, Total Endurance, UNI/O, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A and other countries SQTP is a service mark of Microchip Technology Incorporated in the U.S.A All other trademarks mentioned herein are property of their respective companies © 2007, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved Printed on recycled paper Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified © 2007 Microchip Technology Inc DS01137A-page 19 WORLDWIDE SALES AND SERVICE AMERICAS ASIA/PACIFIC ASIA/PACIFIC EUROPE Corporate Office 2355 West Chandler Blvd Chandler, AZ 85224-6199 Tel: 480-792-7200 Fax: 480-792-7277 Technical Support: http://support.microchip.com Web Address: www.microchip.com Asia Pacific Office Suites 3707-14, 37th Floor Tower 6, The Gateway Harbour City, Kowloon Hong Kong Tel: 852-2401-1200 Fax: 852-2401-3431 India - Bangalore Tel: 91-80-4182-8400 Fax: 91-80-4182-8422 India - New Delhi Tel: 91-11-4160-8631 Fax: 91-11-4160-8632 Austria - Wels Tel: 43-7242-2244-39 Fax: 43-7242-2244-393 Denmark - Copenhagen Tel: 45-4450-2828 Fax: 45-4485-2829 India - Pune Tel: 91-20-2566-1512 Fax: 91-20-2566-1513 France - Paris Tel: 33-1-69-53-63-20 Fax: 33-1-69-30-90-79 Japan - Yokohama Tel: 81-45-471- 6166 Fax: 81-45-471-6122 Germany - Munich Tel: 49-89-627-144-0 Fax: 49-89-627-144-44 Atlanta Duluth, GA Tel: 678-957-9614 Fax: 678-957-1455 Boston Westborough, MA Tel: 774-760-0087 Fax: 774-760-0088 Chicago Itasca, IL Tel: 630-285-0071 Fax: 630-285-0075 Dallas Addison, TX Tel: 972-818-7423 Fax: 972-818-2924 Detroit Farmington Hills, MI Tel: 248-538-2250 Fax: 248-538-2260 Kokomo Kokomo, IN Tel: 765-864-8360 Fax: 765-864-8387 Los Angeles Mission Viejo, CA Tel: 949-462-9523 Fax: 949-462-9608 Santa Clara Santa Clara, CA Tel: 408-961-6444 Fax: 408-961-6445 Toronto Mississauga, Ontario, Canada Tel: 905-673-0699 Fax: 905-673-6509 Australia - Sydney Tel: 61-2-9868-6733 Fax: 61-2-9868-6755 China - Beijing Tel: 86-10-8528-2100 Fax: 86-10-8528-2104 China - Chengdu Tel: 86-28-8665-5511 Fax: 86-28-8665-7889 Korea - Daegu Tel: 82-53-744-4301 Fax: 82-53-744-4302 China - Fuzhou Tel: 86-591-8750-3506 Fax: 86-591-8750-3521 Korea - Seoul Tel: 82-2-554-7200 Fax: 82-2-558-5932 or 82-2-558-5934 China - Hong Kong SAR Tel: 852-2401-1200 Fax: 852-2401-3431 Malaysia - Kuala Lumpur Tel: 60-3-6201-9857 Fax: 60-3-6201-9859 China - Nanjing Tel: 86-25-8473-2460 Fax: 86-25-8473-2470 Malaysia - Penang Tel: 60-4-227-8870 Fax: 60-4-227-4068 China - Qingdao Tel: 86-532-8502-7355 Fax: 86-532-8502-7205 Philippines - Manila Tel: 63-2-634-9065 Fax: 63-2-634-9069 China - Shanghai Tel: 86-21-5407-5533 Fax: 86-21-5407-5066 Singapore Tel: 65-6334-8870 Fax: 65-6334-8850 China - Shenyang Tel: 86-24-2334-2829 Fax: 86-24-2334-2393 Taiwan - Hsin Chu Tel: 886-3-572-9526 Fax: 886-3-572-6459 China - Shenzhen Tel: 86-755-8203-2660 Fax: 86-755-8203-1760 Taiwan - Kaohsiung Tel: 886-7-536-4818 Fax: 886-7-536-4803 China - Shunde Tel: 86-757-2839-5507 Fax: 86-757-2839-5571 Taiwan - Taipei Tel: 886-2-2500-6610 Fax: 886-2-2508-0102 China - Wuhan Tel: 86-27-5980-5300 Fax: 86-27-5980-5118 Thailand - Bangkok Tel: 66-2-694-1351 Fax: 66-2-694-1350 Italy - Milan Tel: 39-0331-742611 Fax: 39-0331-466781 Netherlands - Drunen Tel: 31-416-690399 Fax: 31-416-690340 Spain - Madrid Tel: 34-91-708-08-90 Fax: 34-91-708-08-91 UK - Wokingham Tel: 44-118-921-5869 Fax: 44-118-921-5820 China - Xian Tel: 86-29-8833-7252 Fax: 86-29-8833-7256 10/05/07 DS01137A-page 20 © 2007 Microchip Technology Inc [...]... voltage To estimate size for the coupling capacitor, the capacitor derivative equation can be used dV I C = C × dt The rate of change of voltage across the capacitor is related to the amount of current through the capacitor and the size or energy storage capability of the capacitor For the SEPIC converter coupling capacitor, the voltage is approximated to be a DC value when deriving the duty cycle The. .. connected to VOUT, the anode of the schottky diode is connected to the SEPIC coupling capacitor The voltage across the coupling capacitor voltage is equal to VIN; the voltage across the diode is equal to VOUT - (-VIN) or VOUT + VIN The peak SEPIC diode current occurs when the switch is turned off The peak diode current is equal to the peak current in W2, plus the peak current in W1 or 3.14A The average... applications, the SEPIC diode will block current flow from the battery back to the input The reverse leakage current of the selected schottky diode can be a critical parameter, if low battery drain is desired Low schottky diode forward drop is also a key parameter; the low drop improves converter efficiency The maximum reverse voltage across the SEPIC diode occurs during the switch on time The cathode of the. .. used to determine the size of the inductor necessary High switching frequency has several advantages, smaller ripple current, lower peak and RMS current and lower volt-time product on the inductor core This leads to a small, low- cost solution SEPIC Switch Current and Voltage Calculations The switch current (IQ1) is equal to the combination of the winding currents during the switch on time When the switch... no more than 5% of the voltage across the capacitor or the input voltage In this example, the input voltage and C1 DC voltage is 12V, so there should be no more than 5% or 600 mV of ripple on the coupling capacitor In this example there is an average of 2A flowing through the coupling capacitor during the switch on time The on time is approximately 26% or 520 ns To keep the capacitor voltage ripple... shown in Figure 12 (during the switch on time), the current in winding 2 (output current) is flowing through the coupling capacitor C1 During the switch off time, the C1 current is equal to the current in winding number 1 (W1) As previously discussed, the W1 current is equal to the average input current Therefore, the worst case or maximum RMS current in the coupling capacitor will occur at maximum... voltage As the input voltage decreases, the current in W1 increases During the switch on time, the current flowing in W2 is equal to the current flowing in C1 When the switch turns off, the current quickly changes magnitude and direction so that the current flowing in C1 is equal to the current in W1, magnitude and direction DS01137A-page 17 AN1137 As an approximation, the RMS current in C1: I C1 ( RMS )... Technology Inc DS01137A-page 15 AN1137 Inductor Winding Current Calculation The first step to calculating the inductor winding current is to know the maximum output power For this constant current battery charger application, the output power is simply the maximum output voltage times the charge current P OUT = V OUT × I CHARGE Maximum output voltage is equal to 4.2V (1 battery @ 4.2V) POUT =4.2V x 2... range using a schottky diode for the output rectifier is around 85% • PIN = 8.4 Watts / 0.85 or 9.88 Watts • IIN = PIN / IIN - IIN = +12V / 0.88 Watts - IIN = 1.21 A With IIN and IBATT known, the average inductor current for each winding is known Inductor Ripple Current For the coupled inductor, the effective inductance is twice the value of the inductor, this is a result of 2x the voltage across 2x the. .. Since the value of L is proportional to n2, the effective inductance is twice the actual value of the inductor 2×V -2 n © 2007 Microchip Technology Inc A 10 µH inductor looks like a 20 µH inductor (for coupled inductors only) Larger inductance reduces ripple current and operates in the continuous mode at lighter loads, an advantage over non-coupled inductor solutions The input and output inductor ... Fax: 4 3-7 24 2-2 24 4-3 93 Denmark - Copenhagen Tel: 4 5-4 45 0-2 828 Fax: 4 5-4 48 5-2 829 India - Pune Tel: 9 1-2 0-2 56 6-1 512 Fax: 9 1-2 0-2 56 6-1 513 France - Paris Tel: 3 3-1 -6 9-5 3-6 3-2 0 Fax: 3 3-1 -6 9-3 0-9 0-7 9 Japan... China - Chengdu Tel: 8 6-2 8-8 66 5-5 511 Fax: 8 6-2 8-8 66 5-7 889 Korea - Daegu Tel: 8 2-5 3-7 4 4-4 301 Fax: 8 2-5 3-7 4 4-4 302 China - Fuzhou Tel: 8 6-5 9 1-8 75 0-3 506 Fax: 8 6-5 9 1-8 75 0-3 521 Korea - Seoul Tel: 8 2-2 -5 5 4-7 200... Tel: 88 6-3 -5 7 2-9 526 Fax: 88 6-3 -5 7 2-6 459 China - Shenzhen Tel: 8 6-7 5 5-8 20 3-2 660 Fax: 8 6-7 5 5-8 20 3-1 760 Taiwan - Kaohsiung Tel: 88 6-7 -5 3 6-4 818 Fax: 88 6-7 -5 3 6-4 803 China - Shunde Tel: 8 6-7 5 7-2 83 9-5 507