Lumped Elements for RF and Microwave Circuits phần 7 docx

289 Via Holes depends on the separation between via holes. The sharp increase in equivalent inductance for W = 2 mils at 2 GHz is reported to be due to the numerical precision problem in the analysis. 9.2.5 Measurement-Based Model The measurement-based model of a via hole can be derived using one-port or two-port S-parameters. In this case the via hole structure is represented by a lumped-element EC. Figure 9.11(a) shows the top view of a via hole embedded in the transmission line TRL. The equivalent circuit representation of this element given in Figure 9.11(b) is composed of a series inductance and shunt capacitance associated with the via hole pad and the shunt inductance and Figure 9.11 (a) Via hole embedded in TRL standard and (b) model of a via hole. 290 Lumped Elements for RF and Microwave Circuits resistance of the metal plug. The comparison between the via hole model and its three measured data sets, shown in Figure 9.12, indicates an excellent correlation. Table 9.2 provides model parameters for two pad dimensions and two substrate thicknesses. A via hole model has also been validated by comparing the measured and simulated S 11 data for a 5-pF capacitor terminated by a via hole using a 75- ␮ m- thick GaAs substrate. 9.3 Via Fence Low-cost RF and microwave systems mandate a higher level of integration and more circuit functions in a smaller package. In other words, one needs to integrate RF/microwave circuits, digital circuits, and interconnect and bias lines in a compact package to lower the volume and cost. When such components are placed in proximity to each other, a fraction of the power present on the Figure 9.12 Measured versus modeled input reflection coefficient of a via hole. Substrate thickness = 125 ␮ m. 291 Via Holes Table 9.2 Physical Dimensions and Equivalent Model Parameters Values for Via Hole of Figure 9.11 Physical Dimensions VIA75-1 VIA75-2 VIA125-1 VIA125-2 Units Width, W 175 225 175 225 ␮ m Length, ᐉ 175 225 175 225 ␮ m Substrate thickness, h 75 75 125 125 ␮ m Equivalent Circuit Values VIA75-1 VIA75-2 VIA125-1 VIA125-2 Units Inductance, L 1 0.017 0.023 0.022 0.029 nH Inductance, L 2 0.003 0.003 0.005 0.005 nH Resistance, R 0.02 0.02 0.02 0.02 ⍀ Shunt capacitance, C 0.09 0.13 0.07 0.10 pF main structure is coupled to the secondary structure. The power coupled is a function of the physical dimensions of the structure, TEM (transverse electro- magnetic) or non-TEM, mode of propagation, the frequency of operation, and the direction of propagation of the primary power. In these structures, there is a continuous coupling between the electromagnetic fields, known as parasitic coupling or cross-talk. Such parasitic coupling can take place between the distrib- uted matching elements or closely spaced lumped elements, affecting the electri- cal performance of the circuit in several ways depending on the type of circuit. It may change the frequency response in terms of frequency range and band- width and degrade the gain/insertion loss and its flatness, input and output VSWR, and many other characteristics including output power, power-added efficiency, and noise figure. This coupling can also result in the instability of an amplifier circuit or create feedback resulting in a peak or a dip in the measured gain response or a substantial change in a phase-shifter response. In general, this parasitic coupling is undesirable and an impediment in obtaining an optimal solution in a circuit design. However, this coupling effect can be reduced by using metal-filled via holes known as a via fence [21–23]. Via fences provide an electric wall between the fringing fields and are commonly used in single and multilayer ceramic technologies, silicon and GaAs MIC technologies, and system-on-a package (SOP) technology. In this structure, con- necting via top pads by a strip improves the isolation between the structures by6to10dB. To accurately determine such coupling, an electromagnetic simulator such as three-dimensional finite-element method was used [24]. The results of the analysis is for the structure, fabricated in LTCC technology, are shown in Figure 9.13. The parameters for the structure are given in Table 9.3. 292 Lumped Elements for RF and Microwave Circuits Figure 9.13 Via hole fence: (a) cross-sectional view and (b) four-port circuit configuration. Table 9.3 Summary of Substrate, Microstrip, and Via Hole Parameters Used to Calculate Isolation Between Two Microstrip Lines in the Via Fence Structure Substrate: Glass-ceramic ⑀ r = 5.2 Thickness h = 0.25 mm Microstrip: Width W = 0.414 mm (50⍀) Length ᐉ = 11.7 mm Distance between lines D = 1.814 mm Via: Diameter d = 0.25 mm Distance between microstrip and via fence S = 0.75 mm Figure 9.14 shows the calculated forward coupling between two microstrip lines, with and without a via fence, versus frequency. Here, G is the distance between via posts, center to center, and ‘‘no strip’’ means vias are not connected by the strip on the top side. The data show that the via fence with strip improves coupling by about 8 dB, whereas via posts without strip degrade coupling 293 Via Holes Figure 9.14 Coupling coefficient versus frequency for various G/h values. (From: [24].  2001 IEEE. Reprinted with permission.) at high frequencies. Larger spacing between vias also degrades coupling with frequency. 9.3.1 Coupling Between Via Holes The coupling between two via holes was analyzed using an EM simulator. Figure 9.15(a) shows the structure, where D is the separation between via hole pads. The pad is a square geometry having a side dimension of 165 ␮ m. The substrate is 125- ␮ m-thick GaAs. The coupling between two via holes versus frequency for four separations (15, 100, 200, and 400 ␮ m) is shown in Figure 9.15(b). The coupling for offset via holes, as shown in Figure 9.16(a), was also evaluated. Figure 9.16(b) shows the coupling coefficient versus frequency for four offset S values (40, 80, 165, and 330 ␮ m) and D = 60 ␮ m. The coupling is a strong function of distance between via hole plugs and does not depend on their orientations. 9.3.2 Radiation from Via Ground Plug At low frequencies, a via hole acts as a short; however, as the frequency increases, the reactive component and radiation resistance become significant at high frequencies. Cerri et al. [25] have calculated the radiation resistance using a full-wave analysis. In this case, the via hole is represented by a series combination 294 Lumped Elements for RF and Microwave Circuits Figure 9.15 (a) Two via hole configuration and (b) simulated coupling coefficient versus frequency. of an inductor and a radiation resistance. Figure 9.17 shows a plot of calculated frequency dependence of radiation resistance for an 80- ␮ m-diameter via hole. The GaAs substrate thickness was 200 ␮ m. Although the radiation resistance becomes significant at millimeter-wave frequencies, its value below 20 GHz is negligible. 9.4 Plated Heat Sink Via In MMICs, active devices such as FETs, HEMTs, and HBTs have via hole grounds for source pads and emitter pads, respectively. Such ground connections have appreciable inductance to reduce gain at higher frequencies. To lower source inductance and reduce thermal resistance of FETs, plated heat sinks (PHS) are widely used for discrete devices. In this case (shown in Figure 9.18), each source pad is connected to the PHS through the holes underneath these pads. 9.5 Via Hole Layout When an MMIC chip is mounted on a substrate (alumina, BeO, AlN, and so on), establishing a good ground connection between the back of the chip and 295 Via Holes Figure 9.16 (a) Two via holes in offset configuration and (b) simulated coupling coefficient versus frequency. Figure 9.17 Radiation resistance of a via hole. 296 Lumped Elements for RF and Microwave Circuits Figure 9.18 PHS geometry. the back of the substrate is essential. Here the substrate is epoxied/soldered to a conductor or a fixture. A poorly grounded MMIC chip may exhibit reduced performance or spurious oscillations [26]. To minimize these effects, several via holes are used to connect the mounting pad under the footprint of the chip to case ground. The layout of such via holes and their numbers helps greatly in the elimination of resonant modes in the mounting pad. A large number of via holes, permitted by substrate technology and cost, are generally used to ensure the reproduction of the MMIC performance. Several other factors includ- ing thinner substrates, larger via hole size, via spacings of less than ␭ /20 at the maximum operating frequency, and chips having minimum possible out-of- band gain help in achieving acceptable RF performance and eliminate spurious oscillations. References [1] Ferry, D. K., (Ed.), Gallium Arsenide Technology, Indianapolis, IN: Howard W. Sams, 1985, Chap. 6. [2] Goyal, R., (Ed.), High Frequency Analog Integrated Circuit Design, New York: John Wiley, 1995, Chap. 4. [3] Goldfarb, M. E., and R. A. Pucel, ‘‘Modeling Via Hole Grounds in Microstrip,’’ IEEE Microwave Guided Wave Lett., June 1991, Vol. 1, pp. 135–137. [4] Wang, T., R. F. Harrington, and J. Mautz, ‘‘Quasi-Static Analysis of a Microstrip Via Through a Hole in a Ground Plane,’’ IEEE Trans. Microwave Theory Tech., June 1988, Vol. 36, pp. 1008–1013. [5] Rautio, J. C., and R. F. Harrington, ‘‘An Electromagnetic Time-Harmonic Analysis of Shielded Microstrip Circuits,’’ IEEE Trans. Microwave Theory Tech., August 1987, Vol. MTT-35, pp. 726–730. 297 Via Holes [6] Finch, K. L., and N. G. Alexopoulos, ‘‘Shunt Posts in Microstrip Transmission Lines,’’ IEEE Trans. Microwave Theory Tech., November 1990, Vol. 38, pp. 1585–1594. [7] Maeda S., T. Kashiwa, and I. Fukai, ‘‘Full Wave Analysis of Propagation Characteristics of a Through Hole Using the Finite Difference Time-Domain Method,’’ IEEE Trans. Microwave Theory Tech., December 1991, Vol. MTT-39, pp. 2154–2159. [8] Tsai, W. J., and J. T. Aberle, ‘‘Analysis of a Microstrip Line Terminated With a Shorting Pin,’’ IEEE Trans. Microwave Theory Tech., April 1992, Vol. MTT 40, pp. 645–651. [9] Becker, W. D., P. Harms, and R. Miltra, ‘‘Time Domain Electromagnetic Analysis of a Via in a Multilayer Computer Chip Package,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1992, pp. 1129–1232. [10] Jansen, R. H., ‘‘A Full-Wave Electromagnetic Model of Cylindrical and Conical Via Hole Grounds for Use in Interactive MIC/MMIC Design,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1992, pp. 1233–1236. [11] Sorrentino, R., et al., ‘‘Full Wave Modeling of Via-Hole Grounds in Microstrip by Three Dimensional Mode Matching Technique,’’ IEEE Trans. Microwave Theory Tech., December 1992, Vol. MTT-40, pp. 2228–2234. [12] Visan, S., O. Picon, and V. Fouad Hanna, ‘‘3D Characterization of Air Bridges and Via Holes in Conductor-Backed Coplanar Waveguides for MMIC Applications,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1993, pp. 709–712. [13] Eswarappa, C., and W. J. R. Hoefer, ‘‘Time Domain Analysis of shorting Pins in Microstrip Using 3-D SCN TLM,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1993, pp. 917–920. [14] Cerri, G., M. Mongiardo, and T. Rozzi, ‘‘Full-Wave Equivalent Circuit of Via Hole Grounds in Microstrip,’’ Proc. 23rd European Microwave Conf., 1993, pp. 207–208. [15] Tsai, M. J., et al., ‘‘Multiple Arbitrary Shape Via-Hole and Air-Bridge Transitions in Multi-Layered Structures,’’ IEEE Trans. Microwave Theory Tech., Vol. 44, December 1996, pp. 2504–2511. [16] LaMeres, B. J., and T. S. Kalkur, ‘‘Time Domain Analysis of Printed Circuit Board Via,’’ Microwave J., Vol. 43, November 2000, pp. 76–84. [17] LaMeres, B. J., and T. S. Kalkur, ‘‘The Effect of Ground Vias on Changing Signal Layers in a Multilayered PCB,’’ Microwave Opt. Tech. Lett., Vol. 28, February 2001, pp. 257–260. [18] Sadhir, V. K., I. J. Bahl, and D. A. Willems, ‘‘CAD Compatible Accurate Models of Microwave Passive Lumped Elements for MMIC Applications,’’ Int. J. Microwave Millime- ter-Wave Computer-Aided Engineering, Vol. 4, April 1994, pp. 148–162. [19] Hoffman, R. K., Handbook of Microwave Integrated Circuits, Norwood, MA: Artech House, 1987, Chap. 10. [20] Swanson, D. G., ‘‘Grounding Microstrip Lines with Via Holes,’’ IEEE Trans. Microwave Theory Tech., Vol. 40, August 1992, pp. 1719–1721. [21] Ponchak, G. E., et al., ‘‘The Use of Metal Filled Via Holes for Improving Isolation in LTCC RF and Wireless Multichip Packages,’’ IEEE Trans. Advanced Packaging, Vol. 23, February 2000, pp. 88–99. 298 Lumped Elements for RF and Microwave Circuits [22] Gipprich, J. W., ‘‘EM Modeling of Via Wall Structures for High Isolation Stripline,’’ IEEE MTT-S Int. Microwave Symp. Dig., San Diego, CA, June 1994, pp. 78–114. [23] Gipprich, J., and D. Stevens, ‘‘Isolation Characteristics of Via Structures in High Density Stripline Packages,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1998. [24] Ponchak, G. E., et al., ‘‘Experimental Verification of the Use of Metal Filled Via Hole Fences for Crosstalk Control of Microstrip Lines in LTCC Packages,’’ IEEE Trans. Advanced Packaging, Vol. 24, February 2001, pp. 76–80. [25] Cerri, G., M. Mongiarzdo, and T. Rozzi, ‘‘Radiation from Via-Hole Grounds in Microstrip Lines,’’ IEEE MTT-S Int. Microwave Symp. Dig., 1994, pp. 341–344. [26] Swanson, D., D. Baker, and M. O’Mahoney, ‘‘Connecting MMIC Chips to Ground in a Microstrip Environment,’’ Microwave J., Vol. 36, December 1993, pp. 58–64. [...]... ‘‘High-Q and Low-Loss Matching Network Elements for RF and Microwave Circuits, ’’ IEEE Microwave Magazine, Vol 2, September 2000, pp 64 73 316 Lumped Elements for RF and Microwave Circuits [16] Bessemoulin, A., et al., ‘‘A Simple Airbridge Analytical Model in Coplanar Waveguides for MMIC Applications,’’ Microwave Optical Tech Lett., Vol 17, March 1998, pp 265–2 67 [ 17] Sadhir, V K., I J Bahl, and D A... an 88-␮ mwide line and an 88-␮ m-wide line crossed over a 20-␮ m-wide line (80⍀) Figure 10.13(a) shows the EC model used to represent the airbridge crossover, and Figure 10.13(b) compares the measured and simulated performance for the 20-␮ m-wide line airbridge Table 10.1 lists the model parameter values obtained for the GaAs IC process [ 17] 312 Lumped Elements for RF and Microwave Circuits Figure 10.10... capacitance and inductance per unit length of a multilayer GaAs microstrip of various values of d and W, for h = 125 ␮ m and t = 4.5 ␮ m: (a) capacitance for airbridge, (b) capacitance for crossover, and (c) inductance for crossover 308 Lumped Elements for RF and Microwave Circuits Figure 10.6 Calculated inductance and capacitance per unit length versus airbridge height Figure 10.10 shows the calculated capacitance... Models of Microwave Passive Lumped Elements for MMIC Applications,’’ Int J Microwave and Millimeter-Wave Computer-Aided Engineering, Vol 4, April 1994, pp 148–162 11 Transformers and Baluns Inductor transformers are employed in RF and microwave circuits for various applications including impedance matching, power dividers/combiners, double balanced mixers, power amplifiers, signal coupling, and phase... leakage flux and the shunt inductance is known as the magnetizing inductance These T -equivalent circuits do not show any isolation Figure 11.3 T-equivalent circuit configurations: (a) Z impedances, (b) transformer with M > 0, and (c) transformer with M < 0 324 Lumped Elements for RF and Microwave Circuits between the two input and output ports as desired from a transformer An ideal transformer with... inductor transformers and briefly discuss the various types of such transformers 3 17 318 Lumped Elements for RF and Microwave Circuits 11.1 Basic Theory 11.1.1 Parameters Definition 11.1.1.1 Turns Ratio Transformers are designated by an impedance ratio, for example, 1:n 2 where n is the number of turns ratio In this case, the secondary impedance is n 2 times primary impedance When n > 1, the transformer is... line, and c is the velocity of light If c = 3 × 108 m/s, then C and L are expressed as F/m and H/m, respectively 306 Lumped Elements for RF and Microwave Circuits Figure 10.5 shows the calculated capacitance and inductance per unit length of a microstrip as a function of strip width for various values of air and polyimide thickness under the conductor The substrate was 125-␮ m-thick GaAs (⑀ r = 12.9) and. .. ␮ m, t = 6 ␮ m for (a) Z 0 and (b) ⑀ re C = Cp + Cb Cp = 0.1219ᐉ ͩ √ 29.6 + ln u (10.18a) ͩ ͪͪ 2 1+ u 2 (10.18b) with u = W /t C b = 0.101 ͩ W 1 .78 2 exp − t t ͪ (10.18c) where the dimensions are in microns and capacitances are in femtofarads The inductance L can be evaluated using (2.13) from Chapter 2 with conductor width W and length ᐉ /2 310 Lumped Elements for RF and Microwave Circuits Figure... 4 tanh ( ␤ h 3 ) (10.8b) and Y 1 = ⑀ r1 coth ( ␤ h 1 ) (10.8c) 304 Lumped Elements for RF and Microwave Circuits Figure 10.4 Equivalent transmission-line model Y 4 = ⑀ r4 coth ( ␤ h 4 ) (10.8d) For standard open microstrip, h 2 = h 3 = 0, h 4 = ∞: Y = ⑀ r1 coth ( ␤ h 1 ) + 1 (10.9) For shielded microstrip, h 2 = h 3 = 0 Y = ⑀ r1 coth ( ␤ h 1 ) + ⑀ r4 coth ( ␤ h 4 ) (10.10) For two-layer open microstrip,... of the test structure used for characterizing an airbridge crossover 314 Lumped Elements for RF and Microwave Circuits Figure 10.13 (a) Equivalent circuit model of the airbridge crossover and (b) measured and simulated performance of a 20-␮ m-wide line airbridge Table 10.1 Physical Dimensions and Equivalent Model Values for Two Airbridge Geometries Parameter AIRBRG1 AIRBRG2 Units Width of lower line, . values of d and W, for h = 125 ␮ m and t = 4.5 ␮ m: (a) capacitance for airbridge, (b) capacitance for crossover, and (c) inductance for crossover. 308 Lumped Elements for RF and Microwave Circuits Figure. ( ␤ h 3 ) (10.8b) and Y 1 = ⑀ r1 coth ( ␤ h 1 ) (10.8c) 304 Lumped Elements for RF and Microwave Circuits Figure 10.4 Equivalent transmission-line model. Y 4 = ⑀ r4 coth ( ␤ h 4 ) (10.8d) For standard open. and shunt capacitance associated with the via hole pad and the shunt inductance and Figure 9.11 (a) Via hole embedded in TRL standard and (b) model of a via hole. 290 Lumped Elements for RF and