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Softwar e Radio Arc hitecture: Object-Oriented Approac hes to Wireless Systems Engineering Joseph Mitola III Copyright c !2000 John Wiley & Sons, Inc. ISBNs: 0-471-38492-5 (Hardback); 0-471-21664-X (Electronic) 8 RF/IF Conversion Segment Tradeoffs This chapter introduces the system-level design tradeoffs of the RF conversion segment. Software radios require wideband RF/IF conversion, large dynamic range, and programmable analog signal processing parameters. In addition, a high-quality SDR architecture includes specific measures to mitigate the interference readily generated by SDR operation. I. RF CONVERSION ARCHITECTURES The RF conversion segment of the canonical software radio is illustrated in Figure 8-1. The antenna segment may provide a single element for both trans- mission and reception. In this case, a multicoupler, circulator, or diplexer pro- tects the receiver from the high-power transmission path. In other cases, the transmit and recei ve antennas may be physically separate and may be sepa- rated in frequency. First-generation cellular radio and GSM systems separate downlink and uplink bands by typically 45 MHz to limit interference. The transmission subsystem intersects the RF conversion segment as shown in Figure 8-1. This includes a final stage of up-conversion from an IF, band- pass filtering to suppress adjacent channel interference, and final power am- Figure 8-1 The canonical model characterizes RF/IF se gment interfaces. 265 266 RF/IF CONVERSION SEGMENT TRADEOFFS plification. First-generation cellular systems did not employ power control to any significant degree. CDMA systems, including third-generation (3G) W-CDMA, require power control on each frame (50 to 100 times per sec- ond). SDRs may be implemented with a DAC as the interface between IF up-conversion and the RF segment. Alternatively, a high-speed DAC may d i- rectly feed the final power amplifier. Power amplifiers have less-than-ideal performance, including amplitude ripple and phase distortion. Although these effects may be relatively small, failure to address them may have serious consequences on SDR performance. Amplitude ripple, for example, degrades the transmitted power across the band, particularly near the band edges. IF processing may compensate by preemphasizing the IF signal with the inverse of the power amplifier’s band- edge ripple. Feher [238] describes techniques for compensating a sequence of channel symbols, shaping the transmitted waveform in the time domain to yield better spectral purity in the frequency domain. The concept behind Feher’s patented design is straightforward. Sequential symbols may have the same relative phase, yet the channel-symbol window in which the sinusoids are generated modulates the amplitude at the symbol boundaries. When adja- cent symbols have different phase, this symbol weighting reduces frequency domain sidelobes and hence adjacent-channel interference. Feher suppresses the modulation further with an extended symbol that includes the sequential symbols of the same phase generated with constant amplitude, thus without the weighting-induced amplitude modulation. The result is that energy that normally is redirected into the adjacent channels by the phase discontinu- ities remains within the channel because the discontinuities have been sup- pressed. The receiver subsystem intersection with the RF conversion segment is shown in Figure 8-1 also. This includes the low noise amplifier (LNA), one or more stages of bandpass filtering (BPF), and the translation of the RF to an IF. In conventional radios, a tunable-reference local oscillator (LO) may be shared between the transmitter and receiver subsystems. FH radios of- ten share a fast-tuning LO between the transmitter and receiver. In military applications, the LO executes a frequency-hopping plan defined by a trans- mission security (TRANSEC) module. In commercial systems (e.g., GSM), a fixed frequency-hopping plan that suppresses fades may be used instead of a complex TRANSEC plan. The radio then either transmits or receives on the frequency to which the LO is tuned. Any radio which employs a physi- cally distinct programmable LO may be a programmable digital radio (PDR), a type of SDR, but it is not a software radio. Software radios use lookup tables to define the instantaneous hop frequencies, not physical LOs. This ap- proach, of course, requires a wideband DAC. One advantage of using such a DAC is that the hop frequency settles in the time between DAC samples, typically Wa= 2 : 5—hundreds of nanoseconds. The hop frequency is pure and stable instantly, subject to minor distortions introduced by the final power amplifier. RECEIVER ARCHITECTURES 267 Since the receiver must overcome channel impairments, it may be more complex and technically demanding than the transmitter. Thus, this chapter focuses on receiver design. Again referring to Figure 8-1, IF processing may be null, as may baseband processing. The direct conversion receiver, for example, modulates a reference signal against the received RF (or IF) signal to yield a baseband binary analog waveform in the in-phase and quadrature (I&Q) channels. Although this kind of RF conversion has nonlinear characteristics, it is particularly effective for single-user applications such as handsets. It may not work well for multiuser applications, however. This chapter examines the SDR implications of the RF conversion segment. The following section describes receiver architectures. Programmable compo- nent technology including MEMS and EPACs is described. RF subsystem specifications are then analyzed. The chapter concludes with an assessment of RF/IF conversion architecture tradeoffs. II. RECEIVER ARCHITECTURES This section describes the superheterodyne architecture used in base station applications, the direct con version receiver used in handsets, and related re- search. A. The Superheterodyne Receiver The Watkins-Johnson company [239] publishes the frequency plans of its re- ceivers, an example of which is shown in Figure 8-2. This superheterodyne receiver [240] consists of a preselector and two conversion stages. The prese- lector consists of a matrix of bandpass filters and amplifiers that are switched as defined by the frequency plan for the specific frequency to which the re- cei ver is tuned. Th e preselector filters cascade with a low-pass filter and step attenuator that keep the total power of the signal into the first conversion stage within its linear r ange. Each conversion stage includes one LO and additional filtering and am- plification. The first local oscillator is tuned in relatively coarse steps (e.g., 2.5 MHz in Figure 8-2). The first conversion stage converts the RF to 3733.75 MHz. Higher IF frequencies minimize the physical size of the inductors and capacitors used in the filters. The modulator that converts the RF into the initial IF generates sum and difference frequencies in addition to the desired frequency. The bandpass filter then suppresses these intermodulation products. The low-pass filter further suppresses out-of-band energy. An amplifier and pads with variable gain determine the power into the second conversion stage. The operation of the second stage is similar to the first except that it down- converts the 3733.75 MHz to a standard wideband IF, in this case, 21.4 MHz. In addition, this stage has fine-tuning steps of 1 kHz. 268 RF/IF CONVERSION SEGMENT TRADEOFFS Figure 8-2 Superheterodyne receiver architecture. Figure 8-3 Frequency plan suppresses spectral artifacts. Artifacts must be controlled in the conversion process [241, 242]. In addi- tion to the desired sideband, the conversion process introduces thermal noise, undesired sidebands, and LO leakage into the IF signal as shown in Figure 8-3. Thermal noise is shaped by the cascade of bandpass and low-pass filters. Depending on the RF background environment, thermal noise in the receiver may dominate or thermal-like noise or interference from the environment may dominate the noise power. Superconducting IF filters suppress noiselike interference generated in one cellular half-band from a second, immediately adjacent half-band (e.g., RECEIVER ARCHITECTURES 269 12.5 MHz of active signals). See [243] for superconducting filter test results that show a 30 dB suppression of such noise. Undesired sidebands are al- ways present at some very low level because filtering operations suppress sideband energy but do not completely eliminate it. LO leakage occurs be- cause a modulator acts in some ways as a transmission line with imperfect matching. Consequently, part of the power of the LO is transmitted through the modulator to the output. When the IF is processed digitally, these artifacts can be characterized. Long-term averaging using an FFT, for example, w ill reveal shape of the noise and the degree of suppression of the LO leakage and of the undesired sidebands. When designing a PDR, one is concerned that these artifacts not distort the baseband enough to degrade the output SNR or BER unacceptably. When designing an SDR, none of these artifacts should degrade any of the subscriber channels by more than the degradation of the least significant bit (LSB) of the ADC. To accomplish this, the in-band artifacts need to be as uni- form as possible and the maximum level anywhere in the operating band (e.g., in the cell channels) cannot exceed half of the LSB of the ADC. As shown below, this constraint implies that the ADC, postprocessing algorithms, and RF plan must be designed to mutually support each other. Algorithm design- ers who employ floating-point precision at design time may not be familiar with the noise, spurs, and other analog artifacts of the analog RF circuits that limit useful dynamic range constraints. These effects limit the digital dynamic range, and thus reduce the requirements for arithmetic precision in the digi- tal hardware and software. Thus, the effects of each of the disparate analog, digital, and software-signal processing stages have an effect on the sampled signal. When these effects are properly balanced, the wideband superheterodyne receiver yields hundreds of analog subscriber channels that have been struc- tured for the ADC. A s a result, the id eal software radio base station replaces hundreds of parallel narrowband analog channels with one wideband chan- nel digitized by a wideband ADC, followed by hundreds of parallel digital channels. Since the digital channels inherently cost less than analog channels, the software radio base station may be more cost-effective than the baseband digital design. Yet most base stations deployed up to 1999 had a baseband digital architecture, not an SDR architecture. The inadequacy of the prior generation of ADC technology explains this situation as discussed in the se- quel. Wideband ADCs were within about 6 to 10 dB of the performance required to effectively compete with baseband architectures in the base sta- tion. By June 2000, digital IF base stations began shipping, but manufacturers did not publically disclose this fact in order to protect this competitive advan- tage. Tsurumi’s discussion of zero-IF filtering with up-conversion in a handset architecture provides an innovative approach to multiple-conversion receivers for handsets [244]. By heterodyning multiple bands to zero-IF, Tsurumi pre- filters any of the commercial standards using a simple programmable low-pass 270 RF/IF CONVERSION SEGMENT TRADEOFFS Figure 8-4 Alcatel direct con version receiver. filter. Subsequent up-conversion before digitizing yields a standard digital IF for multiple commercial standards. B. Direct Conversion Receiver The superheterodyne receiver is relatively complex. Its wideband performance is appropriate for base station applications where hundreds of subscriber chan- nels are to be processed at once. But suppose there is only one channel of interest as in the handset receiver. In this case, there is little benefit to the wideband performance of the superheterodyne receiver. Instead, a direct conversion receiver may be more appropriate [245]. The homodyne receiver translates RF to baseband, with the center frequency tuned to zero Hz in one step. The direct conversion receiver is a homodyne receiver that may use nonzero baseband center frequency and may also demodulate the signal into baseband bitstreams in the same circuit. LO leakage and DC bias can be significant problems with such an approach is used for wideband digital signal processing. On the other hand, Alcatel’s direct conversion GSM receiver represents the kind of approach taken in a viable commercial product. It selects channels via switched capacitor filters in a mixed-signal integrated circuit (IC) as shown in Figure 8-4. The RC-CR network generates quadrature phases [246]. The feedback loop at the output of the modulators is filtered for the GSM’s 280 kHz RF channel bandwidth in such a way that the I&Q amplifiers yield level-shift analog baseband signals. This analog signal has two nominal states, corresponding to the two channel symbol states of the MSK waveform. Siemens [247], Philips, and numerous other manufacturers make similar chip sets [248]. See [249] for a direct-conversion GPS receiver. In the past, gallium arsenide (GaAs) circuit technology was necessary for RF circuits, precluding one from implementing the RF circuitry and the mi- crocontroller of a handset with the same circuit technology. Differences in power supply, thermal properties, and bonding between CMOS and GaAs RECEIVER ARCHITECTURES 271 complicated handset design. Recently, howe ver, CMOS silicon RF 50 W to 40 GHz has been reported. One of the .18 micron CMOS chips [250] supports 2.4 GHz RF at 1.8 Volts. CMOS devices that have been demonstrated include low-noise amplifiers, mixers, differential oscillators, IF strips, and RF power amplifiers with 1 W output and 40 to 50% efficiency at 1 to 2 GHz [251]. C. Digital-RF Receivers PhillipsVision [252] created some excitement by announcing a software-radio on a chip. The interesting aspect of their product announcement is that the de- modulator is said to “operate at RF.” Due to the necessarily vague nature of the statements, it is impossible to determine the exact nature of the demodulation process. This announcement plus the recent interest in digital demodulation at RF makes it useful to address this alternative. The comments below may not be representative of the PhillipsVision product, but they reflect research approaches to digital demodulation at RF. Since GHz clocks can be fabricated in single ASICs, one may employ such a clock to demodulate certain modulation types at RF. One approach is the one-bit direct conversion digital receiver, w hich may be called the RF zero-crossing demodulator. With this approach, the RF is amplified until it is hard-limited into a square wave. Reference square waves are synthesized for each channel-symbol state. An MSK waveform, for example, has two square waves. One corresponds to the mark, say, the lower of the two frequencies. By generating digital streams at mark and space frequencies and counting the number of coincidences between mark and space streams in the incoming RF signal, one can estimate the state of the RF waveform. A bit-timing logic state- machine can then determine bit timing to produce the baseband bitstream. All this can be implemented for a single channel-modulation type in an FPGA using less than ten thousand gates. One advantage of this architecture is that the bit patterns for the channel states may be stored in a lookup table. Differ- ent waveforms at different frequencies correspond to different lookup tables. By using clever data-compression techniques, the lookup tables may be kept compact in spite of the large number of entries in the table. A similar approach simply counts zero-crossings of the RF. Once the vari- ance of N zero-crossing counts becomes small, a signal is present. The strong- est of two or more cochannel signals will be reflected in the subsequent counts for CIR > 7 dB. This phenomenon is the digital equivalent of FM capture [245, p. 497]. Random noise generates zero-crossings with large variance, but a sinusoid h as a tight variance. Frequency modulations like GMSK exhibit two different zero-crossing rates, one corresponding to mark, and the other to space. The output of a zero-crossing counter, then, can be gated and reset at the expected channel-symbol rate. A threshold determines whether the channel symbol was mark or space, yielding the baseband bitstream. Timing logic can also estimate and track symbol timing. Although the logic has to operate at the GHz rates of the RF zero-crossings, the counter logic is simplicity itself. 272 RF/IF CONVERSION SEGMENT TRADEOFFS However, certain problems have precluded this receiver architecture from being widely used. First, low-power, high-speed logic has not been available until recently. Thus, the architecture seems timely. In addition, however, the incoming signal cannot be equalized using this type of receiver. The BER floor therefore is worse than that of an equalized receiver. In addition, the recovery of a timing reference is difficult in fading, again raising the BER floor. There does not appear to be much in-depth discussion in the literature of su ch receiver architectures. D. Interference Suppression The first line of defense in suppressing interference is in the antenna and RF conversion segment of the receiver. Physical antenna separation, frequency separation, programmable analog notch filters, and active cancellation are steps that help control interference at RF. In addition, the software of a well- conceived SDR will include mutual constraints among air interfaces that could be invoked simultaneously so that self-generated interference is avoided or minimized. 1. Frequency Separation Interference introduced into a receiver from out-of- band energy created by a nonideal transmitter is the convolution of the out-of- band signal with the bandpass characteristic of the receiver [245]. Although out-of-band interference of high-performance transmitters rolls off to less than " 100 dB within 20 MHz of the transmitted frequency, r adios with less EMI control may present more like " 70 or " 60 dB of rolloff. The presence of a half dozen signals within the overall operating band can then cause substantial interference. FDD standards separate the uplink and downlink to minimize this kind of interference. SDRs operating in TDD bands can create dynamic FDD nets by a protocol that dynamically define uplink, downlink, and frequency separation. This is a novel approach to interference suppression. 2. Programmable F ilters The application of a programmable interference suppression filter is illustrated in Figure 8-5. The filter m ay be called a roof- ing filter b ecause the interference captures the dynamic range, establishing a maximum (roof) and minimum (floor) linearly processable signal level. It is also called a cosite filter in military jargon because the interference may be generated by the colocation of two transmitters in the same locale (site). Prior to the application of the roofing filter, the roof of the dynamic range is so high that weak signals fall below the floor, resulting in dropped calls. After the application of the filte r, the roof has been lowered such t hat the dynamic range is now on the noise floor. Although the interference is still present, it has been suppressed enough to control the available dynamic range. In order for this approach to b e effective, the f ilters have to have low insertion loss, pro- grammable center frequency, and programmable bandwidth. Amplitude and phase ripple across the band has to be kept to near zero to avoid distorting the other subscriber signals. RECEIVER ARCHITECTURES 273 Figure 8-5 Workable situation for roofing filter. Figure 8-6 Roofing filters distort subscriber signals. Not all situations can be addressed effectively using roofing filters, however. If there are more than a fe w strong interference signals in the passband, the roofing filters may introduce excessive distortion into the subscriber signals. This si tuation is illustrated i n Figure 8-6. Factors that determine the number and characteristics of allowed roof- ing filters include the modulation of the subscriber signals, and the band- width of the interference relative to the overall passband. If the subscriber signals are robust to phase and amplitude distortion (e.g., FSK), then more filters or filters that introduce more severe distortion may be used. If the sub- scriber signals are phase-sensitive (e.g., 16 QAM proposed in many of the 3G alternatives), no more than one analog roofing filter is likely to be work- able. 3. Active Cancellation Active cancellation is the process of introducing a replica of the transmitted signal into the receiver so that it may be some- 274 RF/IF CONVERSION SEGMENT TRADEOFFS how subtracted from the input signal. A detailed treatment of cancellation techniques is beyond the scope of this text, but the following introduces the essential notions. Active blanking of radar signals from the input to communications systems on the same platform is an example of active cancellation. In this case, the radar transmitter provides a control line that is active a few microseconds before it transmits so that the communications system can activate a grounding circuit. The RF stage passes no signal at all to the rest of the communications system until the control line is inactive [245]. Active communications cancellation circuits may delay the transmitted sig- nal and attenuate it in such a way that the transmitted and recei ved signals are exactly out of phase, shifted by ¼ radians (at RF or IF) with respect to each other. In principle, such a circuit should cause the transmitted signal to be completely removed from the r eceived s ignal. In p ractice, the cancellation is not ideal. In part, this is due to the inexactness of fabrication of analog circuits. In part, modulation of the transmitted signal distorts each IF sinusoid slightly, and the filtering-induced distortion through the transmitting antenna and into the receiving antenna (or through the circulator) differs slightly from the dis- tortion of the cancellation circuit. The result is that simple linear techniques can achieve only about 10 to 20 dB of cancellation. Complex phase-tracking circuits can improve performance, but nonlinear techniques are required to approach 30 to 40 dB. Few of the nonlinear techniques are in the public do- main. The cancellation that is needed is the difference between the maximum nondistorting input signal and the radiation level that reaches the receiving antenna. Required-Cancellation =( Peak energy at the output of the receiver antenna terminals ) (Maximum linear energy) If this power is not suppressed or dissipated, it will capture the roof of the dynamic range and cause either intermodulation distortion or lost subscribers or both. Not all cancellation has to be accomplished using analog circuits. Any cancellation that occurs in the early stages of RF amplification and filtering also improves system linearity and contributes to dynamic range improvement just like roofing filters. Residual components may be further suppressed using digital techniques. 4. Software-based Interference Mitigation SDR architecture exacerbates interference mitigation by driving the radio platforms toward the use of wideband antennas and RF. It also can contribute to interference suppres- [...]... effects of RFI and EMI RFI originates with high-power radio sources that are external to the software radio and its host platform EMI originates within the host hardware or host platform Otherwise, the two are very similar Consider RFI RF/IF CONVERSION ISSUES 285 Figure 8-13 Critical issues in RF conversion in a military context A SPEAKeasy-like radio may have to operate within a kilometer of a high-power... and IF processing segments of the canonical software -radio architecture? Which of these functions are amenable to enhancement through the introduction of digital techniques? EXERCISES 287 2 Consider a wideband radio that implements frequency hopping with a fast-hopping LO What steps are necessary to migrate that implementation toward a software -radio architecture? What benefits would accrue through... colocated node This distance may be estimated using round-trip leading-edge delay techniques similar to the way radio distance measuring equipment (DME) operates [399] An SDR with a 100 MHz ADC/DAC channel and an FPGA with access to the digital IF signal could send a DME signal to be transponded by nearby radios The internal delays can be calibrated so that the distance can be estimated to within 100 feet or... interference-generation potential In addition, not all mutual constraints are as simple as those of the minimalistic type shown above This notion of mutual constraints among waveform families in the context of some host radio platform is a theme that will be further developed in subsequent chapters as more types of potentially problematic interactions are examined The combinatorial growth of mutual constraints is one... causes unpleasant surprises during the integration process The analysis, testing, and management of such mutual constraints therefore emerges as a central theme of the design and implementation of software radios RF COMPONENT TECHNOLOGY 277 III RF COMPONENT TECHNOLOGY This section provides highlights of RF component technology relevant to the development of SDR platforms One objective is to characterize... vehicles that place a premium on the size, weight, and power consumption of electronic systems, such as tactical aircraft MEMS switches and tunable capacitors were demonstrated in FY98 to function for radio frequencies up to 40 GHz They were to be inserted into antenna interface units for the Comanche Helicopter and the F-22 Fighter, targeting a frequency range of 30 MHz to 400 MHz in FY 00 [262] An... widely used air-bag controller, one of the first examples of MEMS integrated onto a single chip 3 SDR Applications MEMS components enhance the possibilities for the programmability needed by software radios in at least two ways Initially, MEMS switches may select among arrays of analog circuits and components so that the RF conversion segment has more degrees of freedom This is a direct application... push-to-talk (PTT) AM/FM voice, EPLRS, and GSM The entries on the diagonal limit the number of channels that can be used in each mode to less than #mode$ max, where #mode$ is PTT, EPLRS, or GSM If the radio has four channels, it may be capable of supporting all four as push-to-talk channels, but it may have some capacity limit to only one EPLRS channel and only two GSM channels When used in combination,... in RF conversion in a military context A SPEAKeasy-like radio may have to operate within a kilometer of a high-power (90 dBm) troposcatter communications system Metal structures may reflect some of the radio energy with nonlinear intermodulation products that inject appreciable narrowband power levels into the SDR receiver Received power of 0 to "10 dBm narrowband artifacts is not impossible If the narrowband... filters can extend the dynamic range of cellular applications All these developments bode well for the introduction of SDR base stations and for the migration of those products toward the ideal software radio during the next ten to twenty years In handset applications, multimode RF-power ICs increase the flexibility of the handset to support multiple modes The pivotal decisions for RF conversion in the . plan. The radio then either transmits or receives on the frequency to which the LO is tuned. Any radio which employs a physi- cally distinct programmable LO may be a programmable digital radio (PDR), a. translation of the RF to an IF. In conventional radios, a tunable-reference local oscillator (LO) may be shared between the transmitter and receiver subsystems. FH radios of- ten share a fast-tuning LO. programmable LO may be a programmable digital radio (PDR), a type of SDR, but it is not a software radio. Software radios use lookup tables to define the instantaneous hop frequencies, not physical LOs.